Dual-frequency matching circuit

ABSTRACT

The connection topology of input terminals ( 2 ), elements ( 4   a   , 4   b   , 4   c  and  4   d ) and load ( 5 ) is designed similarly to a “seven-segment display” that is often used to display numerals on a calculator or a digital watch. Specifically, suppose in the three horizontally running segments of the seven-segment display, the top and bottom ones are associated with the input terminals ( 2 ) and the load ( 5 ) is allocated to the other horizontal one. Then, the other four vertical segments are associated with the elements ( 4   a   , 4   b   , 4   c  and  4   d ), which may be an inductor with an inductance of 2.521 nH, an inductor with an inductance of 76.157 nH, an inductor with an inductance of 1.907 nH, and a capacitor with a capacitance of 1.429 pF, respectively. By adopting this circuit configuration, the total number of elements can be reduced to four and the loss can be reduced significantly. Since the resonant circuits can be eliminated and the size of the ladder circuit can be reduced, impedance matching is achieved with a high degree of stability in spite of a variation in the impedance of the load ( 5 ).

This is a continuation of International Application No.PCT/JP2008/002353, with an international filing date of Aug. 28, 2008,which claims priority of Japanese Patent Application No. 2007−222413,filed on Aug. 29, 2007, the contents of which are hereby incorporated byreference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a dual-frequency impedance matchingcircuit to be inserted between an antenna and an RF circuit in a mobileterminal in order to carry out impedance matching between the antennaand the RF circuit in two arbitrary frequency bands.

2. Description of the Related Art

As cellphone services have become amazingly popular nowadays, there areincreasing demands for an even higher degree of mobility and even moreversatile telecommunications services. To meet such demands, it is nowone of the major technological objects to develop a mobile terminal thathas an even smaller size and yet can use multiple telecommunicationssystems that are currently operating on mutually different frequencybands (such a device is called a “multi-band device”). Quite the sameobject is shared by an antenna that is an important device operating asa radio wave input/output interface. That is to say, development of aneven smaller antenna operating on multiple different frequency bands(which is called a “multi-band antenna”) is awaited.

In actually developing a mobile terminal, however, it is extremelydifficult to realize good antenna properties on multiple desiredfrequency bands just by optimizing the configuration of the antenna.That is why the final frequency adjustment and good impedance matchingwith an RF circuit often get done by inserting an appropriate matchingcircuit between the antenna and the RF circuit. Currently, the frequencybands utilized by various cellphone services are in 800-900 MHz rangeand 1.5-2 GHz range. To realize a multi-band mobile terminal, itsantenna should operate on both of these two frequency bands. However,these two frequency bands are so far apart from each other that it isdifficult for a normal single-frequency matching circuit to carry outflexible matching and adjustment on both of these two frequency bands.That is why to achieve the object described above, it is preferable toapply a dual-frequency matching circuit that can carry out independentmatching on those two frequency bands.

In such a background, some conventional dual-frequency matching circuitsthat have been adopted so far include a ladder circuit consisting ofmultiple single-frequency matching circuits and multiple resonantcircuits (see Japanese Patent Application Laid-Open Publication No.2004-242269 (page 18 and FIG. 1) and Japanese Patent ApplicationLaid-Open Publication No. 2006-325153 (page 14 and FIG. 1), forexample). FIG. 11 is a circuit block diagram illustrating a circuitconfiguration for a conventional dual-frequency matching circuitdisclosed in Japanese Patent Application Laid-Open Publication No.2004-242269 (page 18 and FIG. 1).

In FIG. 11, the frequency dependency of impedance (or a single-terminalS parameter) at an output terminal 102 is already known and a load 101corresponds to an antenna in the situation described above. And the load101 is connected to a power supply 107 by way of a conventionaldual-frequency matching circuit 108 consisting of first, second andthird matching circuits 103, 104 and 105. As shown in the block diagramsin FIG. 11, these matching circuits 103, 104 and 105 are parallel orserial resonant circuits, each of which is made up of inductors andcapacitors.

The conventional dual-frequency matching circuit 108 shown in FIG. 11operates as an impedance transformer such that in two desired frequencybands, the circuit 108 transforms the impedance of the load 101 at theoutput terminal 102 into the impedance value of the power supply 107 atthe input terminal 106. That is why in those two frequency bands, thepower supplied from the power supply 107 can be passed to the load 101efficiently without experiencing reflection attenuation.

Consider each of these three matching circuits 103, 104 and 105 as asingle circuit block. In that case, the conventional dual-frequencymatching circuit 108 shown in FIG. 11 is composed of the two fundamentaltypes of single-frequency matching circuits 121 a and 121 b shown inFIG. 12 (see Robert E. Collin, —An IEEE Press ClassicReissue—Foundations for Microwave Engineering (Second Edition, IEEEPress Series on Electromagnetic Wave Theory), A John Wiley & Sons, Inc.,Publication, ISBN 0-7803−6031-1 (page 323, FIG. 5.17) (hereinafter,Non-Patent Document No. 1), for example), and they are coupled togetherso as to form a ladder circuit 131 as shown in FIG. 13. FIG. 12illustrates circuit block diagrams showing the circuit configurations ofthe two fundamental types of single-frequency matching circuitsdisclosed in Non-Patent Document No. 1 and FIG. 13 is a circuit blockdiagram illustrating the circuit configuration of a ladder circuit foruse in a conventional dual-frequency matching circuit. It should benoted that the ladder circuit 131 is a circuit configuration that isordinarily used in various types of filters.

The function of the conventional dual-frequency matching circuit 108 isequivalent to transmitting an RF signal from the input terminal 106 tothe load 101 on two desired frequency bands without causing anyreflection attenuation. That is why by adopting the ladder circuit 131shown in FIG. 13, designing a dual-frequency matching circuit isequivalent to designing a band-pass filter, of which the pass bands arethose two desired frequency bands. Consequently, in designing theconventional dual-frequency matching circuit 108, the conventionalfilter designing method can be used effectively, and matching with theinput terminal 106 can be done relatively flexibly on two desiredfrequency bands without depending on the frequency response of theimpedance at the load 101. These are advantages of the conventionalconfiguration.

However, the conventional configuration has the following two drawbacks.

Firstly, it is difficult to reduce the loss caused by the dual-frequencymatching circuit. To improve the quality of cellphone services, thetransmission and receiving properties of mobile terminals must beimproved. The transmission and receiving properties are improved mainlyby reducing the transmission loss between the antenna and the RFcircuit. That is why the loss to be caused by the dual-frequencymatching circuit inserted there is preferably as little as possible. Theconventional configuration, however, needs too many elements (includinginductors and capacitors) and must use a number of resonant circuits,and therefore, is a problem as far as loss reduction is concerned.

Another problem is that it is difficult to stabilize the matchingproperty with respect to the variation in the impedance of the load 101.Normally, when a mobile terminal is used, the user's hand or head comesclose to the antenna. That is why the frequency dependency of theimpedance for the antenna is affected by how the device is used. Forthat reason, to ensure stabilized transmission and receiving quality,the matching property must be stabilized with respect to the variationin the impedance of the antenna. However, since the conventional circuitdescribed above includes a lot of resonant circuits, of which theelectrical properties (represented by a two-terminal S parameter) varysteeply with the frequency, the matching property is easily affected bythe variation in the impedance of the load 101. Furthermore, in theladder circuit 131, the impedance is transformed in eachsingle-frequency matching circuit 121 a, 121 b (see FIG. 12) and theladder circuit is composed of a number of such single-frequency matchingcircuits. That is why the ladder circuit itself is sensitive to thevariation in the impedance of the load 101. In view of theseconsiderations, the conventional configuration described above is stillto be improved in terms of stability, too.

In order to overcome the problems described above, the present inventionhas an object of providing a dual-frequency matching circuit that causeslittle loss and that achieves high stability with respect to a variationin the impedance of a load.

SUMMARY OF THE INVENTION

A dual-frequency matching circuit according to the present inventionincludes: first and second input terminals that receive a first RFsignal with a frequency of 0.85 GHz and a second RF signal with afrequency of 2.05 GHz, respectively, from an RF circuit with animpedance of 50Ω; first and second output terminals that are connectedto an antenna; and a group of circuit elements that are connectedbetween the input terminals and the output terminals. The group ofcircuit elements includes first, second, third and fourth elements. Thefirst and fourth elements are connected in series between the first andsecond input terminals and the second and third elements are connectedin series between the first and second input terminals. The first outputterminal is connected to a connection node between the first and fourthelements. The second output terminal is connected to a connection nodebetween the second and third elements. The first element is an inductorwith an inductance of 2.521 nH. The second element is an inductor withan inductance of 76.157 nH. The third element is an inductor with aninductance of 1.907 nH. And the fourth element is a capacitor with acapacitance of 1.429 pF.

In one preferred embodiment, the impedance of the antenna is 54.4−8.3i Ω(where i is an imaginary unit) at a frequency of 0.85 GHz and 50.4+8.2iΩ (where i is an imaginary unit) at a frequency of 2.05 GHz,respectively.

In a specific preferred embodiment, the antenna is an inverted F antennato be built in a cellphone.

The dual-frequency matching circuit of the present invention can resolvethe two major technological issues (i.e., loss reduction andstabilization of matching property).

Other features, elements, processes, steps, characteristics andadvantages of the present invention will become more apparent from thefollowing detailed description of preferred embodiments of the presentinvention with reference to the attached drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit block diagram illustrating a circuit configurationfor a dual-frequency matching circuit as a first preferred embodiment ofthe present invention.

FIG. 2 shows sign and other notations to illustrate how to determineelement constants in the dual-frequency matching circuit of the firstpreferred embodiment of the present invention.

FIG. 3A is a circuit diagram illustrating how to expand an inductor(that is a single constituent element of the dual-frequency matchingcircuit of the first preferred embodiment of the present invention) intoa circuit consisting of multiple inductors, and FIG. 3B is a circuitdiagram illustrating how to expand a capacitor (that is a singleconstituent element of the dual-frequency matching circuit of the firstpreferred embodiment of the present invention) into a circuit consistingof multiple capacitors.

FIG. 4 illustrates perspective view showing the dimensions of ananalytical model for a mobile terminal with an antenna as a specificexample of the first preferred embodiment of the present invention.Specifically, FIG. 4A is a perspective view showing the overalldimensions of the analytical model, and FIG. 4B is a perspective viewshowing the detailed dimensions of the antenna portion thereof.

FIG. 5 shows the frequency dependency of the radio frequency propertiesof the analytical model shown in FIG. 4 at the output terminals 3 withan impedance of 50Ω in the specific example of the first preferredembodiment of the present invention. Specifically, FIG. 5A shows thefrequency dependency of a voltage standing wave ratio and FIG. 5B is aSmith chart of the single-terminal S parameters.

FIG. 6 is a table of element constants in the dual-frequency matchingcircuit of the present invention that was designed for the analyticalmodel shown in FIG. 4 in the specific example of the first preferredembodiment of the present invention.

FIG. 7 is a perspective view showing the dimensions and position of amodeled hand that was added to the analytical model shown in FIG. 4 in aspecific example of the first preferred embodiment of the presentinvention.

FIG. 8 is a table showing the change ratios of the impedance matchingbandwidth of the dual-frequency matching circuits of the presentinvention, which were designed as shown in FIG. 6, in this specificexample of the first preferred embodiment of the present invention, whena hand came close to the mobile terminal as shown in FIG. 7.

FIG. 9 shows block diagrams of conventional dual-frequency matchingcircuits. Specifically, FIG. 9A is a block diagram of a dual-frequencymatching circuit based on the single-frequency matching circuitdisclosed in Non-Patent Document No. 1 shown in FIG. 12A. On the otherhand, FIG. 9B shows a dual-frequency matching circuit based on thesingle-frequency matching circuit disclosed in Non-Patent Document No. 1shown in FIG. 12B.

FIG. 10A shows that there are no solutions for the elementconfigurations, element constants of the conventional dual-frequencymatching circuits and bandwidth change ratios of the matching bands whena hand comes close to them, and FIG. 10B is a table showing the elementconfigurations, element constants of the conventional dual-frequencymatching circuits and bandwidth change ratios of the matching bands whena hand comes close to the mobile terminal. Specifically, FIG. 10B showsa table that was calculated for the circuit blocks shown in FIG. 9B.

FIG. 11 is a circuit block diagram illustrating a circuit configurationfor a conventional dual-frequency matching circuit.

FIGS. 12A and 12B illustrate circuit block diagrams showing the circuitconfigurations of the two fundamental types of single-frequency matchingcircuits disclosed in Non-Patent Document No. 1.

FIG. 13 is a circuit block diagram illustrating the circuitconfiguration of a ladder circuit for use in a conventionaldual-frequency matching circuit.

FIG. 14 is a perspective view showing the detailed dimensions of anantenna portion according to a second specific example of the firstpreferred embodiment of the present invention (with antenna resonantfrequencies of 0.85 GHz and 1.55 GHz).

FIG. 15 is a perspective view showing the detailed dimensions of anantenna portion according to the second specific example of the firstpreferred embodiment of the present invention (with antenna resonantfrequencies of 0.85 GHz and 1.7 GHz).

FIG. 16 is a perspective view showing the detailed dimensions of anantenna portion according to the second specific example of the firstpreferred embodiment of the present invention (with antenna resonantfrequencies of 0.85 GHz and 2.05 GHz).

FIG. 17A shows the frequency dependency of the radio frequencyproperties of the analytical model shown in FIG. 14 at the outputterminals 3 with an impedance of 50Ω in the second specific example ofthe first preferred embodiment of the present invention. Specifically,FIG. 17A shows the frequency dependency of a voltage standing wave ratioand FIG. 17B is a Smith chart of the single-terminal S parameters.

FIG. 18A shows the frequency dependency of the radio frequencyproperties of the analytical model shown in FIG. 15 at the outputterminals 3 with an impedance of 50Ω in the second specific example ofthe first preferred embodiment of the present invention. Specifically,FIG. 18A shows the frequency dependency of a voltage standing wave ratioand FIG. 18B is a Smith chart of the single-terminal S parameters.

FIG. 19A shows the frequency dependency of the radio frequencyproperties of the analytical model shown in FIG. 16 at the outputterminals 3 with an impedance of 50Ω in the second specific example ofthe first preferred embodiment of the present invention. Specifically,FIG. 19A shows the frequency dependency of a voltage standing wave ratioand FIG. 19B is a Smith chart of the single-terminal S parameters.

FIG. 20 is a table of element configuration and element constants in thedual-frequency matching circuit of the present invention that wasdesigned for the analytical model shown in FIG. 14 in the secondspecific example of the first preferred embodiment of the presentinvention.

FIG. 21 is a table of element configuration and element constants in thedual-frequency matching circuit of the present invention that wasdesigned for the analytical model shown in FIG. 15 in the secondspecific example of the first preferred embodiment of the presentinvention.

FIG. 22 is a table of element configuration and element constants in thedual-frequency matching circuit of the present invention that wasdesigned for the analytical model shown in FIG. 16 in the secondspecific example of the first preferred embodiment of the presentinvention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

Hereinafter, preferred embodiments of the present invention will bedescribed with reference to the accompanying drawings.

Embodiment

FIG. 1 is a circuit block diagram illustrating a circuit configurationfor a dual-frequency matching circuit as a preferred embodiment of thepresent invention. As shown in FIG. 1, the dual-frequency matchingcircuit 1 of this preferred embodiment includes input terminals 2consisting of first and second input terminals 2 a and 2 b and outputterminals 3 consisting of first and second output terminals 3 a and 3 b.An RF circuit (not shown) is connected to the input terminals 2, while aload 5 is connected to the output terminals 3.

The dual-frequency matching circuit 1 of this preferred embodimentincludes four elements 4 a, 4 b, 4 c and 4 d, which are lumped constantelements and each of which is either an inductor or a capacitor. Thetypes of these elements 4 a, 4 b, 4 c and 4 d, which should be eitherinductors or capacitors, and the specific values of their respectiveelement constants are determined unequivocally by the impedance value ofthe load 5 that has been defined in advance on the two frequency bands,where matching should be achieved, and by the impedance value of the RFcircuit connected to the input terminals 2. As for specifically how todetermine them, it will be described in detail later.

The connection structure of the input terminals 2, the elements 4 a, 4b, 4 c and 4 d and the load 5 is similar to that of a so-called“seven-segment display”, which is often used to display numerals on acalculator or a digital watch. More specifically, suppose in the threehorizontally running segments of the seven-segment display, the top andbottom segments are associated with the input terminals 2 and the load 5is allocated to the other horizontally running segment. Then, theremaining four vertical segments are associated with the elements 4 a, 4b, 4 c and 4 d.

In this case, the dual-frequency matching circuit 1 of the presentinvention has a geometrically symmetric circuit configuration. That iswhy even if the elements 4 a, 4 b, 4 c and 4 d are rearranged in any ofthe following manners, the rearranged circuit will exhibit quite thesame characteristic as the original one. A first one of thoserearrangements is to exchange the positions of the elements 4 a and 4 band the positions of the elements 4 d and 4 c with each other at thesame time. This first rearrangement is equivalent to moving the inputterminals 2 that are located on the left-hand side of the dual-frequencymatching circuit 1 of the present invention to the right-hand side asshown in FIG. 1. A second one of those rearrangements is to exchange thepositions of the elements 4 a and 4 d and the positions of the elements4 b and 4 c with each other at the same time. This second rearrangementis equivalent to changing the connections of the two output terminals ofan external circuit (not shown in FIG. 1). For example, if the outputterminals of the external circuit not shown in FIG. 1 form an unbalancedwire circuit, the rearrangement means changing the terminals to beconnected to the ground plane of the output terminal of the externalcircuit not shown in FIG. 1 between the upper and lower ones of theinput terminals 2 shown in FIG. 1. It can also be confirmed by Equations(2) of the design process to be described later that the first andsecond rearrangements produce an electrically equivalent circuit. Thatis why even though there may be a number of circuit configurationsavailable for the dual-frequency matching circuit 1 of the presentinvention according to the design process to be described later, thecircuit configurations to be correlated with each other in the first andsecond rearrangements are not independent of each other but may becombined into a single circuit configuration.

Next, it will be described specifically how to determine the elementconstant values of the elements 4 a, 4 b, 4 c and 4 d. Since each ofthese elements 4 a, 4 b, 4 c and 4 d is either an inductor or acapacitor, the impedance of each element is a pure imaginary number.Thus, in the following description, the impedances of the respectiveelements are supposed to have the signs shown in FIG. 2.

FIG. 2 shows sign and other notations to illustrate how to determineelement constants in the dual-frequency matching circuit of thispreferred embodiment of the present invention. In FIG. 2, the smallletter “i” represents an imaginary unit. That is to say, i=(−1)̂(½). Theimpedance value Z0 of the RF circuit connected to the input terminals 2is a real number value and is usually 50Ω. On the other hand, theimpedance of the load 5 is normally a complex quantity, which hasfrequency dependency and the real part Zr(f) and the imaginary partZi(f) (where f denotes the frequency) of which are represented by tworeal number quantities.

As described above, the impedance of each element is represented by areal number quantity αj(f) (where j=1, 2, 3 or 4). Also, depending onwhether each element is an inductor or a capacitor, αj(f) (where j=1, 2,3 or 4) is defined by the following Equation (1):

${\alpha_{j}(f)} = \left\{ {\begin{matrix}{- \frac{1}{\left( {2\pi \; f} \right)C_{j}}} & ({capacitor}) \\{\left( {2\pi \; f} \right)L_{j}} & ({inductor})\end{matrix},\left( {{j = 1},2,3,4} \right)} \right.$

In Equation (1), Lj and Cj are element constants (i.e., the inductanceand capacitance, respectively) of the j^(th) element. At this point intime, those specific values have not been determined yet and are stillunknown numbers. That is why the specific Lj and Cj values are obtainedby solving the following four simultaneous equations (which will becollectively referred to herein as Equations (2)) at the two frequenciesf1 and f2 at which the impedance matching should be achieved.

$\begin{matrix}{{{{A\left( f_{k} \right)} - {Z_{0}{C\left( f_{k} \right)}}} = 0}{{{{B\left( f_{k} \right)} - {Z_{0}{D\left( f_{k} \right)}}} = 0},\left( {{k = 1},2} \right)}\left\{ \begin{matrix}{{A(f)} = {{Z_{r}(f)}\left\{ {{\alpha_{2}(f)} + {\alpha_{3}(f)}} \right\} \left\{ {{\alpha_{1}(f)} + {\alpha_{4}(f)}} \right\}}} \\{{B(f)} = {{{Z_{i}(f)}\left\{ {{\alpha_{2}(f)} + {\alpha_{3}(f)}} \right\} \left\{ {{\alpha_{1}(f)} + {\alpha_{4}(f)}} \right\}} +}} \\\left\lbrack {{\left\{ {{\alpha_{2}(f)} + {\alpha_{3}(f)}} \right\} {\alpha_{1}(f)}} + {\alpha_{4}(f)} +} \right. \\\left. {{\alpha_{2}(f)}{\alpha_{3}(f)}\left\{ {{\alpha_{1}(f)} + {\alpha_{4}(f)}} \right\}} \right\rbrack \\{{C(f)} = {{{Z_{i}(f)}\left\{ {{\alpha_{1}(f)} + {\alpha_{2}(f)} + {\alpha_{3}(f)} + {\alpha_{4}(f)}} \right\}} +}} \\{\left\{ {{\alpha_{1}(f)} + {\alpha_{2}(f)}} \right\} \left\{ {{\alpha_{3}(f)} + {\alpha_{4}(f)}} \right\}} \\{{D(f)} = {{- {Z_{r}(f)}}\left\{ {{\alpha_{1}(f)} + {\alpha_{2}(f)} + {\alpha_{3}(f)} + {\alpha_{4}(f)}} \right\}}}\end{matrix} \right.} & (2)\end{matrix}$

Equations (2) may be solved in the following manner. First, each of theelements 4 a, 4 b, 4 c and 4 d is provisionally supposed to be acapacitor or an inductor. Then, αj(f) (where j=1, 2, 3 or 4) becomes afunction relative to the frequency f, including four undeterminedelement constants (Lj or Cj), according to Equation (1). Then, theimpedance Zr(f) and Zi(f) of the load 5 and the impedance value Z0 ofthe RF circuit connected to the input terminals 2, of which thefrequency characteristics are already known, and αj(f) (where j=1, 2, 3or 4), of which the specific function form has been determined byEquation (1), are substituted into the right sides of the third throughsixth ones (as counted from the top) of Equations (2), therebyconstituting A(f), B(f), C(f) and D(f). Then, A(f), B(f), C(f) and D(f),of which the specific function forms relative to the frequency f are nowknown, are substituted into the two-condition equation on the first andsecond rows of Equations (2) and two desired frequencies fk (where k=1and 2) are given, thereby obtaining four mutually independent equationswith respect to the four undetermined element constants (Lj or Cj).Then, those four equations are solved simultaneously to derive the fourundetermined element constants. It should be noted that since the numberof undetermined constants agrees with that of independent equations,Equations (2) always have solutions. However, the element constants mustbe positive real numbers. That is why only if positive real numbers areobtained as solutions Lj or Cj, the dual-frequency matching circuit ofthis preferred embodiment can be actually implemented as the circuitshown in FIG. 2.

Sixteen (2̂4=16) different combinations of capacitors and inductors canbe allocated to the elements 4 a, 4 b, 4 c and 4 d. That is why bysolving Equations (2) just as described above for each and every one ofthose sixteen combinations, all possible circuit configurations thatcould be implemented as real circuits can be extracted. After that, itis necessary to perform the processing step of selecting an independentcircuit properly by combining together, into a single circuit, a numberof mutually correlated circuit configurations to be obtained by firstand second rearrangements due to the geometric symmetry of the circuits.And if one of those possible independent circuit configurations thatsatisfies most perfectly the requirements imposed on the antenna asneeded is selected according to the situation, the design process of thedual-frequency matching circuit of this preferred embodiment iscompleted.

Examples of those requirements to be imposed as needed include whetherthe bandwidth that would achieve good matching is sufficiently broad ornot, whether the dual-frequency matching circuit is made up of elementswith small element constants or not (i.e., whether or not there is anyinductor with a large element constant), and whether or not the matchingproperty is affected easily by a variation in the impedance of theantenna. If the dual-frequency matching circuit of the present inventionis designed as a matching circuit for an antenna in a mobile terminal asdescribed above, the last requirement is particularly important.

According to such a configuration, a dual-frequency matching circuit ismade up of four lumped elements, each of which is either a capacitor oran inductor, thereby reducing the minimum required number of elements tofour. And as those elements are coupled together with a circuitconfiguration other than a ladder circuit consisting of resonantcircuits, a dual-frequency matching circuit, which would cause littleloss and would operate with good stability without having its impedancematching easily affected by a variation in the impedance of the load 5,can be provided.

In the preferred embodiment described above, each of the elements 4 a, 4b, 4 c and 4 d is supposed to be either a single inductor or a singlecapacitor. However, if any of those elements is implemented as aninductor, the inductor could be replaced with two or more inductors thatare connected in series together as shown in FIG. 3A, which is a circuitdiagram illustrating how to expand an inductor (that is a singleconstituent element of the dual-frequency matching circuit of thispreferred embodiment) into a circuit consisting of multiple inductors.Likewise, if any of those elements is implemented as a capacitor, thecapacitor could be replaced with two or more capacitors that areconnected in parallel with each other as shown in FIG. 3B, which is acircuit diagram illustrating how to expand a capacitor (that is a singleconstituent element of the dual-frequency matching circuit of the firstpreferred embodiment of the present invention) into a circuit consistingof multiple capacitors. In both of these two cases, however, the overallinductance of the entire circuit or the overall capacitance of theentire circuit needs to agree with the element constant that has beenobtained for a single element by the design process described above.

Specific Example 1

Hereinafter, a specific example of a dual-frequency matching circuitaccording to the present invention will be described. The basicconfiguration of this specific example is the same as the configurationof the preferred embodiment shown in FIG. 1.

FIG. 4 is a perspective view showing the dimensions of an analyticalmodel for a mobile terminal with an antenna as a specific example of thepresent invention. Specifically, FIG. 4A is a perspective view showingthe overall dimensions of the analytical model and FIG. 4B is aperspective view showing the detailed dimensions of the antenna portionthereof. In FIG. 4, the entire analytical model is made of a metallicplate with a thickness of 100 μm and a conductivity of 4.9×10̂7 Sie/m.

As shown in FIG. 4A, the antenna 6 is an inverted F antenna formed byfolding the metallic plate and is connected to the top end of a metallicbox with dimensions of 40 mm×85 mm×5 mm, which represents a model of thehousing 7 of a mobile terminal. The output terminal 3 for inputting anRF signal to the antenna 6 (more specifically, the first output terminal3 a connected to the elements 4 b and 4 c shown in FIG. 1) is located atthe encircled position in FIG. 4A. On the other hand, the second outputterminal 3 b that is short-circuited with the second input terminal 2 bin FIG. 1 corresponds to the grounded housing shown in FIG. 4A.

Also, supposing this analytical model was put in a free space (i.e., aninfinite vacuum), the model was subjected to a radio frequency analysisusing an electromagnetic field simulator IE3D version 11.23, therebyextracting the frequency behaviors of the impedance of the antenna 6,including the influence of the mobile terminal's housing 7 at the outputterminals 3. In this specific example, the antenna had an impedance of103.5−90.1i Ω (where i is an imaginary unit) at a frequency of 0.85 GHzand an impedance of 44.9−15.9i Ω (where i is an imaginary unit) at afrequency of 1.86 GHz, respectively.

Hereinafter, it will be described how to design the dual-frequencymatching circuit of this specific example, which will be connected tothe output terminals 3 of the mobile terminal shown in FIG. 4, by thedesign process of the preferred embodiment described above. As acomparative example, it will also be described how to design adual-frequency matching circuit with the conventional configurationdescribed above. And their degrees of stability with respect to thevariation in the impedance of the antenna 6, including the influence ofthe mobile terminal's housing 7, will be compared to each other, therebyconfirming the superiority of the dual-frequency matching circuit of thepresent invention over the conventional one.

First, the single-terminal S parameters, which were calculated at theoutput terminals 3 of the mobile terminal shown in FIG. 4 by performingelectromagnetic field simulations, are shown in FIG. 5. Specifically,FIG. 5 shows the frequency dependency of the radio frequency propertiesof the analytical model shown in FIG. 4 at the output terminals 3 withan impedance of 50Ω in this specific example. More specifically, FIG. 5Ashows the frequency dependency of a voltage standing wave ratio and FIG.5B is a Smith chart of the single-terminal S parameters.

In this specific example, the two frequencies f1 and f2, at which theimpedance matching should be achieved, are supposed to be 0.85 GHz and1.86 GHz, respectively, and the impedance value of the RF circuit to bematched is supposed to be 50Ω (i.e., Z0=50). As can be seen from FIG.5A, the antenna produces resonances in the vicinity of those frequenciesf1 and f2. However, at the frequency f1, in particular, a sufficientdegree of matching (i.e., voltage standing wave ratio≦2) is notachieved. Such a result is also observed in FIG. 5B. Specifically, asindicated by the solid triangles ▴ in FIG. 5B, at the frequency f1,matching with 50Ω is not realized. That is why the dual-frequencymatching circuit of the present invention is designed and connected tothe output terminals 3, thereby achieving complete matching(corresponding to a voltage standing wave ratio of one) at both of thosetwo frequencies.

The element constants that were calculated by the design processdescribed above are shown in FIG. 6, which is a table of elementconstants in the dual-frequency matching circuit of the presentinvention that was designed for the analytical model of this specificexample shown in FIG. 4. In FIG. 6, the letters C and L in the ElementConfiguration portion of the table denote a capacitor and an inductor,respectively. Also, in FIG. 6, specific element constant values of theelements, which were calculated for the results of electromagnetic fieldsimulations shown in FIG. 5, are shown in the Element Constant portionof the table.

Thus, the dual-frequency matching circuit of this specific exampleincludes: first and second input terminals 2 a, 2 b that receive a firstRF signal with a frequency of 0.85 GHz and a second RF signal with afrequency of 1.86 GHz, respectively, from an RF circuit with animpedance of 50Ω; first and second output terminals 3 a, 3 b that areconnected to an antenna (load 5); and a group of circuit elements thatare connected between the input terminals 2 and the output terminals 3.

The group of circuit elements includes first, second, third and fourthelements 4 a, 4 b, 4 c and 4 d. The first and fourth elements 4 a and 4d are connected in series between the first and second input terminals 2a and 2 b and the second and third elements 4 b and 4 c are connected inseries between the first and second input terminals 2 a and 2 b. Thefirst output terminal 3 a is connected to a connection node between thefirst and fourth elements 4 a and 4 d, and the second output terminal 3b is connected to a connection node between the second and thirdelements 4 b and 4 c.

The group of circuit elements is one of the following two sets:

The first set includes an inductor with an inductance of 12.084 nH, aninductor with an inductance of 5.452 nH, an inductor with an inductanceof 14.508 nH, and a capacitor with a capacitance of 1.934 pF as thefirst, second, third and fourth elements, respectively.

And the second set includes a capacitor with a capacitance of 1.023 pF,an inductor with an inductance of 5.772 nH, a capacitor with acapacitance of 0.904 pF, and an inductor with an inductance of 14.927 nHas the first, second, third and fourth elements, respectively.

When used, every mobile terminal always comes close to the user's handor head. And the degrees of the closeness change according to how he orshe uses it or who uses it. That is why to provide goodtelecommunication quality, it is important to stabilize the matchingproperty with respect to a variation in the impedance of the antennathat will be caused when the user's hand or head comes close to theterminal. Thus the present inventors calculated how much thecharacteristic of the analytical model shown in FIG. 4 deteriorated whena modeled hand 8 came close to that model.

FIG. 7 is a perspective view showing the dimensions and position of amodeled hand that was added to the analytical model shown in FIG. 4 in aspecific example of the first preferred embodiment of the presentinvention. In FIG. 7, the modeled hand 8 was supposed to be a dielectricblock with a uniform dielectric constant of 50 and a uniform dielectricloss of 0.45. The present inventors carried out similar electromagneticfield simulations under these circumstances. The degrees of decline ofthe relative bands that were calculated by those simulations are shownin FIG. 8.

FIG. 8 is a table showing the change ratios of the impedance matchingbandwidth of the dual-frequency matching circuits of the presentinvention, which were designed as shown in FIG. 6, in this specificexample of the first preferred embodiment of the present invention, whena hand came close to the mobile terminal as shown in FIG. 7. In FIG. 8,the “bandwidth change ratio” was calculated by {(bandwidth withhand)−(bandwidth without hand)}/(bandwidth without hand)×100.

As used herein, the “band” is defined to be a frequency band with avoltage standing wave ratio of 2 or less. It can be seen from FIG. 8that the circuit configuration that resulted in the smallest variationeven when a hand came close to the circuit was Case 2.

Even within the scope of the conventional technology described above, adual-frequency matching circuit can also be made up of the same numberof elements (i.e., four elements) as the dual-frequency matching circuitof the present invention. Such a circuit has a circuit configuration inwhich the single-frequency matching circuits with the configurationshown in FIG. 12 are connected together to form a ladder circuit asshown in FIG. 13. And the circuit can be implemented as one of the twoindependent circuits shown in FIG. 9.

FIG. 9 shows block diagrams of conventional dual-frequency matchingcircuits. Specifically, FIG. 9A is a block diagram of a dual-frequencymatching circuit based on the single-frequency matching circuitdisclosed in Non-Patent Document No. 1 shown in FIG. 12A. On the otherhand, FIG. 9B shows a dual-frequency matching circuit based on thesingle-frequency matching circuit disclosed in Non-Patent Document No. 1shown in FIG. 12B. As for these conventional dual-frequency matchingcircuits, the bandwidth change ratios of the matching bands when a handcomes close to the circuits can also be calculated by the same procedureas the one for obtaining the results shown in FIG. 8. The results areshown in FIGS. 10A and 10B.

FIG. 10 shows a table showing the element configurations, elementconstants of the conventional dual-frequency matching circuits andbandwidth change ratios of the matching bands when a hand comes close tothem. Specifically, FIG. 10A shows the results (no solutions) obtainedfor the dual-frequency matching circuit shown in FIG. 9A. FIG. 10B showsa table that was calculated for the dual-frequency matching circuitshown in FIG. 9B. As shown in FIG. 10A, in this specific example, nomatching can be achieved by the dual-frequency matching circuit shown inFIG. 9A. That is why the only circuit configuration available for theconventional dual-frequency matching circuit is the one shown in FIG.9B.

Comparing FIGS. 8 and 10 to each other, it can be seen that the elementconfiguration in Case 2 of the dual-frequency matching circuit of thepresent invention exhibited the highest degree of stability at both ofthe two frequencies (and particularly at the frequency f1). That is tosay, it can be seen that the present invention is superior to the priorart in terms of the ability to ensure stabilized matching property,which is one of the most important requirements for a mobile terminal.

Specific Example 2

Hereinafter, another specific example of a dual-frequency matchingcircuit according to the present invention will be described.

The frequency bands that are already utilized currently and the onesthat will be exploited in the near future can be roughly classified intothe three frequency bands, namely, a low frequency band in a 0.45 GHzrange, an intermediate frequency band in 0.8 GHz, 0.85 GHz and 0.9 GHzranges, and a radio frequency band in 1.5 GHz, 1.7 GHz, 1.8 GHz, 1.9 GHzand 2.0 GHz ranges. Among other things, the intermediate and radiofrequency bands are in particularly high demand for common use. For thatreason, in this specific example, a representative frequency of 0.85 GHzis selected from the intermediate frequency band and threerepresentative frequencies of 1.55 GHz, 1.7 GHz and 2.05 GHz areselected from the radio frequency band to design an antenna first.

These frequencies are selected because all the other frequency bands areasymptotic to one of the four selectable frequencies including 1.86 GHzand because as far as the frequency dependency of the antenna'simpedance is concerned, there should be no significant deviation fromthe designed value at the selected frequency.

In this specific example, the impedance of the antenna is 54.4−8.3i Ω ata frequency of 0.85 GHz, 0.48+46.3i Ω at a frequency of 1.55 GHz,1.80+57.7i Ω at a frequency of 1.7 GHz, and 50.4+8.15i Ω at a frequencyof 2.05 GHz, respectively.

The configuration of this specific example is basically the same as, butis slightly different from, that of the preferred embodiment shown inFIG. 1. One of the differences between the two lies in that thestructure and dimensions of the antenna may change in this specificexample according to the selected frequency in the radio frequency band.In addition, as will be described later, the impedances of the first,second, third and fourth elements that form the dual-frequency matchingcircuit are also different.

The antennas that can be connected to the dual-frequency matchingcircuit of this specific example are shown in FIGS. 14, 15 and 16, whichare perspective views illustrating the detailed dimensions of theantennas of this specific example. The antenna shown in FIG. 14 isdesigned so as to produce resonances at frequencies of 0.85 GHz and 1.55GHz. The antenna shown in FIG. 15 is designed so as to produceresonances at frequencies of 0.85 GHz and 1.7 GHz. And the antenna shownin FIG. 16 is designed so as to produce resonances at frequencies of0.85 GHz and 2.05 GHz. It should be noted that the housing to be loadedwith the antenna shown in FIG. 14, 15 or 16 is supposed to have the sameconfiguration and dimensions as the ones shown in FIG. 4A. Also, in thisspecific example, the antenna and the housing are all made of a metallicplate with a thickness of 100 μm and a conductivity of 4.9×10̂7 Sie/m asin the first specific example described above. The antenna was alsodesigned using an electromagnetic field simulator IE3D version 11.23 asin the specific example described above.

The frequency dependencies of the radio frequency properties at theoutput terminal 3 that is connected to the antenna shown in FIG. 14, 15or 16 in this specific example are shown in FIGS. 17, 18 and 19,respectively. FIGS. 17A, 18A and 19A are graphs each showing thefrequency dependency of a voltage standing wave ratio and FIGS. 17B, 18Band 19B are Smith charts of the single-terminal S parameters.

As can be seen easily from FIGS. 17, 18 and 19, to realize goodimpedance matching between the antenna and the circuit at the twoselected frequencies (i.e., designed frequencies), the dual-frequencymatching circuit needs to be inserted into any of these antennas.

The element configurations and element constants of the dual-frequencymatching circuits of this specific example to be connected to theantennas shown in FIGS. 14, 15 and 16, respectively, are shown in FIGS.20, 21 and 22, which are tables of element constants in thedual-frequency matching circuits that were designed for the analyticalmodels shown in FIGS. 14, 15 and 16, respectively. The specific elementconstants shown in FIGS. 20, 21 and 22 were calculated for the resultsof the electromagnetic field simulations shown in FIGS. 14, 15 and 16.In FIGS. 20, 21 and 22, the letters C and L in the Element Configurationportion of the tables denote a capacitor and an inductor, respectively.Also, in FIGS. 20, 21 and 22, specific element constant values of theelements, which were calculated for the results of electromagnetic fieldsimulations shown in FIG. 5, are shown in the Element Constant portionof the tables.

The dual-frequency matching circuits defined by FIGS. 20, 21 and 22 havethe following configurations:

Configuration Shown in FIG. 20

This dual-frequency matching circuit includes: first and second inputterminals 2 a, 2 b that receive a first RF signal with a frequency of0.85 GHz and a second RF signal with a frequency of 1.55 GHz,respectively, from an RF circuit with an impedance of 50Ω; first andsecond output terminals 3 a, 3 b that are connected to an antenna (load5); and a group of circuit elements that are connected between the inputterminals 2 and the output terminals 3.

The group of circuit elements includes first, second, third and fourthelements 4 a, 4 b, 4 c and 4 d. The first and fourth elements 4 a and 4d are connected in series between the first and second input terminals 2a and 2 b and the second and third elements 4 b and 4 c are connected inseries between the first and second input terminals 2 a and 2 b. Thefirst output terminal 3 a is connected to a connection node between thefirst and fourth elements 4 a and 4 d, and the second output terminal 3b is connected to a connection node between the second and thirdelements 4 b and 4 c.

The first element 4 a is an inductor with an inductance of 4.030 nH. Thesecond element 4 b is an inductor with an inductance of 11.208 nH. Thethird element 4 c is an inductor with an inductance of 2.497 nH. And thefourth element 4 d is a capacitor with a capacitance of 2.233 pF.

Configuration Shown in FIG. 21

This dual-frequency matching circuit includes: first and second inputterminals 2 a, 2 b that receive a first RF signal with a frequency of0.85 GHz and a second RF signal with a frequency of 1.7 GHz,respectively, from an RF circuit with an impedance of 50Ω; first andsecond output terminals 3 a, 3 b that are connected to an antenna (load5); and a group of circuit elements that are connected between the inputterminals 2 and the output terminals 3.

The group of circuit elements includes first, second, third and fourthelements 4 a, 4 b, 4 c and 4 d. The first and fourth elements 4 a and 4d are connected in series between the first and second input terminals 2a and 2 b and the second and third elements 4 b and 4 c are connected inseries between the first and second input terminals 2 a and 2 b. Thefirst output terminal 3 a is connected to a connection node between thefirst and fourth elements 4 a and 4 d, and the second output terminal 3b is connected to a connection node between the second and thirdelements 4 b and 4 c.

The first element 4 a is an inductor with an inductance of 2.132 nH. Thesecond element 4 b is an inductor with an inductance of 8.266 nH. Thethird element 4 c is an inductor with an inductance of 0.596 nH. And thefourth element 4 d is a capacitor with a capacitance of 2.097 pF.

Configuration Shown in FIG. 22

This dual-frequency matching circuit includes: first and second inputterminals 2 a, 2 b that receive a first RF signal with a frequency of0.85 GHz and a second RF signal with a frequency of 2.05 GHz,respectively, from an RF circuit with an impedance of 50Ω; first andsecond output terminals 3 a, 3 b that are connected to an antenna (load5); and a group of circuit elements that are connected between the inputterminals 2 and the output terminals 3.

The group of circuit elements includes first, second, third and fourthelements 4 a, 4 b, 4 c and 4 d. The first and fourth elements 4 a and 4d are connected in series between the first and second input terminals 2a and 2 b and the second and third elements 4 b and 4 c are connected inseries between the first and second input terminals 2 a and 2 b. Thefirst output terminal 3 a is connected to a connection node between thefirst and fourth elements 4 a and 4 d, and the second output terminal 3b is connected to a connection node between the second and thirdelements 4 b and 4 c.

The first element 4 a is an inductor with an inductance of 2.521 nH. Thesecond element 4 b is an inductor with an inductance of 76.157 nH. Thethird element 4 c is an inductor with an inductance of 1.907 nH. And thefourth element 4 d is a capacitor with a capacitance of 1.429 pF.

When the circuit is designed, a number of element configurations otherthan the ones shown in FIGS. 20, 21 and 22 should be obtained assolutions. However, as already described for the first specific example,an element configuration with a large relative band needs to be selectedto minimize the deterioration of the transmission and receivingproperties when a cellphone is used. In addition, as also can be seenfrom FIGS. 8 and 10, the higher the frequency band, the moresignificantly the band changes when a portion of a human body comesclose to the mobile terminal. For that reason, a circuit configurationthat will have its relative band expanded in the radio frequency band ispreferably selected in conclusion.

From these viewpoints, the circuit configurations shown in FIGS. 20, 21and 22 were extracted from the lot of solutions derived.

It should be noted that if the impedance of an antenna for use in thepresent invention varied at each frequency due to the change of thestructures or dimensions of the antenna, the element constant valuesshown in FIGS. 20 to 22 could also change. However, if two frequenciesat which the antenna operates are given, then the structure anddimensions of the antenna that can be used in the present invention aresubstantially determined. For that reason, the structure and dimensionsof the antenna that can be used in practice according to the presentinvention never deviate significantly from the structure and dimensionsof the antenna selected according to the frequency and shown in one ofFIGS. 14 to 16. As a result, the impedances of the antenna at the twofrequencies can also be close to the values described above.

Even if an antenna that has a different structure or differentdimensions from the one shown in FIG. 14, 15 or 16 is used, the elementconstants to be figured out will never deviate significantly from thevalues shown in FIGS. 20 to 22 unless a big difference is producedbetween the impedances of the antenna at the two frequencies describedabove. For example, even if the impedances of the antenna at the twofrequencies have changed to a certain degree due to a variation in thedimensions of the antenna, the dual-frequency matching circuit with theelement constants shown in FIGS. 20 to 22 can still achieve the effectsof the present invention sufficiently.

Stated otherwise, in a situation where the impedance of the antenna isequal to the value of the specific example described above, even if therespective values of element constants do not exactly match the onesshown in FIGS. 20 to 22, the effects of the present invention can stillbe achieved. For example, even if the respective values of the elementconstants have varied by about 50% from the ones shown in FIG. 6, theeffects of the present invention can still be achieved.

A dual-frequency matching circuit according to the present invention ismade up of as small as four elements, and therefore, causes little lossand achieves high stability with respect to a variation in the impedanceof a load. That is why the dual-frequency matching circuit of thepresent invention can be used effectively in an amplifier or a mixer,for example. The present invention is also applicable to a tuned circuitfor use in a plasma generation source for a thin-film deposition systemthat deposits a thin film on a substrate by a physical or chemicalprocess and to a tuned circuit for a magnetron for use in a microwaveoven for heating with microwaves.

While the present invention has been described with respect to preferredembodiments thereof, it will be apparent to those skilled in the artthat the disclosed invention may be modified in numerous ways and mayassume many embodiments other than those specifically described above.Accordingly, it is intended by the appended claims to cover allmodifications of the invention that fall within the true spirit andscope of the invention.

1. A dual-frequency matching circuit comprising: first and second inputterminals that receive a first RF signal with a frequency of 0.85 GHzand a second RF signal with a frequency of 2.05 GHz, respectively, froman RF circuit with an impedance of 50Ω; first and second outputterminals that are connected to an antenna; and a group of circuitelements that are connected between the input terminals and the outputterminals, wherein the group of circuit elements includes first, second,third and fourth elements, and wherein the first and fourth elements areconnected in series between the first and second input terminals and thesecond and third elements are connected in series between the first andsecond input terminals, wherein the first output terminal is connectedto a connection node between the first and fourth elements, and whereinthe second output terminal is connected to a connection node between thesecond and third elements, and wherein the first element is an inductorwith an inductance of 2.521 nH, and wherein the second element is aninductor with an inductance of 76.157 nH, and wherein the third elementis an inductor with an inductance of 1.907 nH, and wherein the fourthelement is a capacitor with a capacitance of 1.429 pF.
 2. Thedual-frequency matching circuit of claim 1, wherein the impedance of theantenna is 54.4−8.3i Ω (where i is an imaginary unit) at a frequency of0.85 GHz and 50.4+8.2i Ω (where i is an imaginary unit) at a frequencyof 2.05 GHz, respectively.
 3. The dual-frequency matching circuit ofclaim 2, wherein the antenna is an inverted F antenna to be built in acellphone.